Electro-optical modulators modulate electrical signals onto a light beam in order to generate a modulated optical beam that carries data. Optical waveguide modulators are well known in the art and are used in a variety of applications.
An electro-optical phase modulator according to the present disclosure may be useful in the prior art 100 G CFP modulator illustrated in FIG. 1. The 100 Gb/s small form factor device 100 consists of an input fiber 10 for receiving an optical signal 11 connected to a lithium niobate substrate 12 having a waveguide 14 with a y-branch 16 to split the optical signal onto two arms, each of which is modulated by phase modulators 18 having an I input electrode 20 and a Q input electrode 22. The two waveguide arms are then combined with a micro-optic polarization multiplexer 24 and output through output fiber 26. Each phase modulator 18 modulates a four amplitude level RF signal that is then polarization multiplexed to result in a dual polarization Quadrature Phase Shift Keying (QPSK) optical signal 28.
The length of the electrodes 20, 22 are adjusted such that the strength of the electro-optic (e/o) response from the Q inputs is ½ that of the e/o response from the I inputs. In FIG. 1, the I RF signal produces a 0° or 180° optical phase shift while the Q RF signal produces a 45° or −45° optical phase shift in addition to that produced by the I RF signal. For the case where RF loss can be neglected in the RF electrodes, the scaling can be accomplished by scaling the electrode lengths by a 2:1 ratio. Hence, the I electrodes 20 use ⅔ of the available total electrode length while the Q electrodes 22 use ⅓ of available length. Each phase modulator 18 is driven by a two-level 25 Gb/sec binary data signal (not illustrated) that is precoded to preserve the digital sequence of the data at the receiver. For example, digital exclusive OR prescaling may be applied to the Q RF input signal but not the I RF input signal, in order to preserve the digital sequence of I and Q at the receiver.
Precoding the RF input signal or signals combined with electro-optic scaling and summation is a much simpler and more compact method to create a 4-level optical signal from two conventional 2-level 25 Gb/sec binary data signals than performing Digital Signal Processing (which consumes significant electrical power) or RF signal summation. In RF signal summation, the two RF electrical signals are summed by brute force where the amplitude of one signal is reduced by 2× relative to the other. RF signal summation requires RF hardware that occupies some space on transmitter card and requires careful control of any timing skew from the two combined binary signals.
FIG. 2 illustrates the optical phase constellation diagram for the QPSK signal produced by each phase modulator 18, where the x and y axis represent real and imaginary components of the optical signal. The 4 RF amplitude levels result in the 4 optical phase states shown in FIG. 2.
FIG. 3 illustrates an example prior art phase modulator 18 comprising two staggered electrodes reversed in order from FIG. 1. For sake of simplicity, ground lines are not shown in any of the Figures; however, there are typically one or two ground lines for each signal electrode creating a ground-signal or ground-signal-ground electrode geometry, respectively. In addition, it is assumed that a signal electrode does not modulate any waveguides once it shifts or jogs off to the side distal from the waveguide.
The electrode design illustrated in FIG. 3 is useful for the case where the RF inputs are co-located near the optical input end of the chip. The two electrodes 20, 22 run parallel to one another from the input end to the output end of the waveguide. The optical signal is first modulated by signal electrode 22. The two electrodes jog over laterally relative to the optical path to allow the other electrode 20 to modulate the waveguide. An electro-optic (e/o) interaction length is defined to be the portion of the electrode where the electrode is in close proximity to the waveguide, such that field lines emanating from the RF signal electrode overlap the optical field in the waveguide. The region of the substrate in which an interaction length modulates the waveguide is also referred to as an interaction region. The lengths, L1 and L2, correspond to the lengths of the e/o interaction region for the two electrodes. Normalized by the total length of interaction regions for both electrodes, L1=0.333 and L2=0.667. The ratio of those lengths, L2/L1=R, is equal to 2. The I data signal is applied to RF electrode 20, while a precoded Q data signal is applied to electrode 22. The order of the electrodes I and Q is reversed from that shown in FIG. 1 in an attempt to reduce RF Loss experienced in electrode 22.
FIG. 4 illustrates the calculated S21 (e/o) frequency response for the I and Q inputs of a phase modulator 18 according to FIG. 3 for the case where the expected RF loss for the electrodes is introduced. The total e/o interaction length is 43 mm. Loss coefficient for RF loss will be defined later; however, both electrodes 20, 22 have approximately 6 dB of electrical loss along their length. Note that the interaction length for electrode 20 is twice that of electrode 22. In addition, electrode 20 has length L1 before electro-optic interaction begins for that electrode. The net effect is that the RF loss in electrode 20 during or before electro-optic interaction is higher than for electrode 22, hence bandwidth for electrode 20, the I signal, is lower than for electrode 22, the Q signal. The thick line in FIG. 4 is the S21 (e/o) response for the I RF signals, where as the thin line is the S21 (e/o) response for the Q RF signals shifted up by 6 dB to account for the 2:1 scaling ratio. Ideally after this shifting, the two curves would lie directly on top of one another providing perfectly matched S21 e/o frequency responses. As can readily be seen in FIG. 4, the S21 (e/o) frequency responses are markedly different, by as much as about 2 dB at 20 GHz, and more at higher frequencies. The 3 dB bandwidth for the I input is about 30 GHz, however the 3 dB bandwidth for the Q input exceeds 50 GHz.
The mismatch in frequency response will degrade the quality of the constellation diagram in FIG. 2 because the I and Q e/o responses will not match for different digital data bit sequences having different electrical spectral content. The dots for the 4 different phase states will become smeared out along the circles illustrated in FIG. 2. For digital sequences dominated by low frequency content, the scaling will be close to 2:1. For digital sequences dominated by high frequency content, the scaling will be less than 2:1. At high frequencies, the optical phase shift produced by the I signal will be significantly less than 180° while the optical phase shift produced by the Q signal will be closer to its nominal value of 45° or −45°.
The phase modulator design can be improved by adjusting the electrode lengths to cause matching of scaled responses at some specific frequency greater than zero, however, the pattern dependent scaling and frequency mismatching of I and Q will still persist. Furthermore, if velocity mismatch (Nrf_error) is considered, the performance of the phase modulators 18 illustrated in FIGS. 1 and 3 decreases further because the separation between the S21 frequency responses increases. An Nrf_error value of 0.03 is within the distribution of typical values for a high volume modulator manufacture line.
For high frequency applications, which are becoming more prevalent, the phase modulators 18 illustrated in FIGS. 1 and 3 do not produce well matched s21 (e/o) frequency responses. It would be advantageous to provide an electro-optical phase modulator having well matched S21 (e/o) frequency responses over a broad range of frequencies.
Some prior art Mach-Zehnder modulators, such as taught in U.S. Patent Publication No. 2011/0158576 published Jun. 30, 2011 to Kissa, U.S. Pat. No. 7,701,630 issued Apr. 20, 2010 to Kissa et al., U.S. Pat. No. 6,501,867 issued Dec. 31, 2002 to Gates, II et al., and U.S. Pat. No. 7,058,241 issued Jun. 6, 2006 to Sugiyama et al., which are each incorporated herein by reference, teach various modulator structures applying domain inversion regions in the waveguide substrate to modulate two waveguides with a single signal electrode with balanced push and pull modulation. They do not teach how to modulator light within a waveguide using more than one signal electrode, nor do they teach how to match the frequency responses corresponding to different signal electrodes.
U.S. Pat. No. 7,277,603 issued Oct. 2, 2007 to Roberts et al. and incorporated herein by reference, teaches digital-to-analog conversion using modulators integrated with electronic drivers.
U.S. Pat. No. 8,044,835 issued October 25, 211 to Ehrlichman and incorporated by reference herein describes a MZ modulator having phase modulators having electrode lengths based on powers of two. Further optimization of electrode lengths could remove small non-linearities in output; but for more significantly non-linear devices, optimizing binary control vectors for a digital-to-analog converter without changing electrode lengths was considered more suitable.
Paragraph 52 of U.S. Pat. No. 8,050,555 issued Nov. 1, 2011 to McBrien and incorporated herein by reference, describes an M-ary Phase Shift Keying (MPSK) modulator using a series of independent phase modulators, each having different lengths according to a binary progression to perform the conversion from binary digital signals to the multi-level optical PSK signal.
In U.S. Pat. No. 6,580,840 issued Jun. 17, 2003 to McBrien et al. and incorporated herein by reference, a prior art electro-optic device increases modulation efficiency by compensating for velocity mismatch between the optical and electrical signals propagating through the device by using phase reversal sections that are co-linear with the optical waveguide; however, it was found that such devices have relatively low modulation efficiency per unit length.
Some other prior art references related to binary weighted phase shift keying modulators include [1] T. Sakamoto, “Coherent synthesis of optical multilevel signals by electrooptic digital-to-analog conversion using multiparallel modulator,” IEEE Journal of Selected Topics in Quantum Electronics, Vol. 16, No. 5, September/October 2010, pp. 1140-1149; [2] Y. Ehrlichman, et. al., “Photonic digital-to-analog conversion and digitally driven integrated optics modulator,” 2011 IEEE International Conference on Microwaves, Communications, Antennas and Electronics Systems (COMCAS), Issue Date: 7-9 Nov. 2011, pp. 1-4; and [3] X. Zhou, et. al., “8×114 Gb/s, 25-GHz-spaced, PolMux-RZ-8PSK transmission over 640 km of SSMF employing digital coherent detection and EDFA-only amplification,” Proceedings from postdeadline papers, 2008 Optical Fiber Communication conference, PDP1, each of which is incorporated herein by reference. [1] teaches optical digital-to-analog conversion performed using parallel modulator topologies. [2] describe optical digital-to-analog conversion performed using serial modulator topologies. [3] describes 8PSK modulation format using cascaded Mach-Zehnder (MZ) interferometer modulators with two phase modulators.